1. Field of the Invention
The present invention relates generally to a mobile communication system, and in particular, to an apparatus and method for detecting a power ratio between a traffic channel and a pilot channel in a mobile communication system for high-speed data transmission.
2. Description of the Related Art
Mobile communication systems have evolved from a conventional communication system that supports a voice-centered service into an advanced communication system that supports a high-capacity data-centered service such as a data service and a multimedia service. Thus, the mobile communication system is evolving into a high-speed, high-quality packet communication system in order to enable high-capacity data transmission. For example, high speed downlink packet access (hereinafter referred to as “HSDPA”) proposed in 3rd Generation Partnership Project (3GPP), standard for 3rd generation asynchronous mobile communication system, or Enhanced Variable-Data Voice (1xEV-DV) proposed in 3rd Generation Partnership Project 2 (3GPP2), standard for 3rd generation synchronous mobile communication system, is a technology proposed for high-capacity, high-speed data transmission. The HSDPA technology is the general term for a high speed downlink shared channel (hereinafter referred to as “HS-DSCH”) which is a downlink data channel for supporting high speed downlink packet data transmission, its associated control channels, for an apparatus, system and method in a wideband code division multiple access (hereinafter referred to as “W-CDMA”) communication system.
In a high-speed packet data transmission system, a technique for adjusting a modulation scheme and a coding scheme according to a radio channel condition has been proposed to transmit high-speed data within a limited frequency band. Particularly, adaptive modulation and coding (hereinafter referred to as “AMC”) has recently been proposed for HSDPA. The AMC technique will now be described herein blow.
AMC refers to a data transmission technique in which a modulation scheme and a coding scheme are determined according to a channel condition between a cell, or Node B, and a user equipment (hereinafter referred to as “UE”). Thus, AMC improves the utilization efficiency of the cell. The AMC includes a plurality of modulation schemes and a plurality of coding schemes, and modulates and codes a channel signal by combining the modulation schemes with the coding schemes. Specifically, each combination of the modulation schemes and the coding schemes is called a modulation and coding scheme (hereinafter referred to as “MCS”), and a plurality of MCSs with level #1 to level #N can be defined according to the number of the MCSs. The AMC technique adaptively determines an MCS level according to a channel condition between a UE and a Node B in which the Node B is wirelessly connected to the UE, thereby improving the entire system efficiency of the Node B. In AMC, Quadrature Phase Shift Keying (QPSK), 8-ary Phase Shift Keying (8PSK) and 16-ary Quadrature Amplitude Modulation (16QAM) are considered for the modulation scheme, and various coding rates of ¼ to 1 are considered for the coding scheme. Although the following description will be made with reference to an asynchronous HSDPA communication system for the convenience of explanation, the following description can also be applied to other mobile communication systems for high-speed data transmission.
When AMC is applied, for UEs having a good channel condition such as the UEs being located in the vicinity of a Node B, i.e., UEs using channels having good quality, a high-order modulation scheme, for example, 8PSK and 16QAM, and a high coding rate are used. In contrast, for UEs located in a remote distance from the Node B, UEs having a poor channel condition such as UEs existing in a cell boundary position, and UEs using channels having poor quality, a low-order modulation scheme, for example, QPSK, and a low coding rate are used. In the case of low-order modulation schemes, especially QPSK used in a conventional 3rd generation mobile communication system, one symbol is located in each quadrant on its constellation, so channel compensation can be performed with only phase estimation. However, in the case of a high-order modulation scheme such as 8PSK or 16QAM, a plurality of symbols are located in each quadrant on its constellation, and a plurality of symbols having different amplitudes can be located in the same phase, so precise estimation for not only phase but also amplitude is required for channel compensation.
As stated above, the modulation scheme and the coding rate applied when a high-speed, high-quality service is provided in a mobile communication system are adaptively used according to a radio channel environment. In addition, when data is transmitted by applying a high-order modulation scheme and a low coding rate, a primary factor of reducing reception of the transmitted data generally occurs in a channel environment between a Node B and a UE. The channel environment that reduces data reception includes an additive white Gaussian noise (hereinafter referred to as “AWGN”), a variation in power of a reception signal due to fading, a Doppler effect due to movement of a UE and a variation in moving speed of the UE, and interference caused by other UEs and multipath signals. Since an original transmission signal is distorted according to a radio channel environment before being received at a receiver, there is a demand for an apparatus for compensating for the distortion in the received signal so that it resembles the transmitted signal. The apparatus is called a “channel estimator.”
In HSDPA, a Node B transmits a common pilot channel (CPICH) signal for channel estimation so that all UEs can receive the CPICH signal. Even in 1xEV-DV for a synchronous system, a base station (BS) transmits a pilot channel (PICH) signal for channel estimation so that all mobile stations (MSs) can receive the PICH signal. In the following description, since both the common pilot channel signal and the pilot channel signal are used for channel estimation, they will be commonly referred to as a “pilot channel,” for purposes of simplicity. The pilot channel is set up between the Node B and the UEs to transmit a pilot signal, and a reception side, or a UE, estimates a channel condition, especially a channel fading phenomenon, between the Node B and the UE by receiving the pilot channel signal. The estimated channel fading is used in restoring a received signal distorted due to a fading phenomenon back to an original signal which was transmitted by the transmission side. Also, the estimated channel fading is used in estimating a power ratio between a traffic channel and a pilot channel (traffic vs. pilot channel power ratio).
The power ratio estimation between a traffic channel and a pilot channel is a necessary procedure for demodulating a signal modulated in a high-order modulation scheme such as 16QAM and 64QAM. If information on the power ratio between a traffic channel and a pilot channel is provided from a transmission side, or a Node B, to a reception side, or a UE, there is no necessity to estimate the power ratio between a traffic channel and a pilot channel. However, a high-speed packet transmission system employing 1xEV-DV or HSDPA in which a high-order modulation scheme of 16QAM or higher order is used is designed so that the estimation should be performed in the reception side in order to remove a signaling load. A method of estimating a power ratio between a traffic channel and a pilot channel at the reception side called “blind power ratio detection” can be used instead of the method of providing information on the power ratio between a traffic channel and a pilot channel from the transmission side to the reception side through signaling. However, primary factors of reducing the blind power ratio detection occurs at the reception side, and the primary factors of reducing the blind power ratio detection are roughly classified into three factors: channel noise, fading phenomenon, and unequal average power.
The unequal average power will now be described with reference to FIG. 1.
FIG. 1 is a graph illustrating an example of a general constellation for 16QAM. Referring to FIG. 1, when a high-order modulation scheme such as 16QAM is applied, respective symbols have different power levels. For example, power of 4 inner symbols being adjacent to a coordinate (0,0) on the constellation becomes Pin=2A2, power of 8 middle symbols on the constellation becomes Pmiddle=10A2, and power of 4 outer symbols on the constellation becomes Pouter=18A2. Thus, the total average power of the 4 inner symbols, the 8 middle symbols and the 4 outer symbols becomes
            P      total        =                                        2            ⁢                          A              2                                +                      10            ⁢                          A              2                                +                      18            ⁢                          A              2                                      3            =              10        ⁢                  A          2                      ,and if A=0.3162, the total average power Ptotal becomes 1. In the following description, it will be assumed that A=0.3162, a particular symbol is represented by Si, and power of the corresponding symbol is represented by <Si>. Here, i is an identifier for identifying a data channel and a pilot channel. If i=d, the i indicates a data channel, while if i=p, the i indicates a pilot channel. For example, <Sd> represents power of a corresponding symbol on a data channel.
The data symbols are transmitted over a traffic channel, and the traffic channel is transmitted together with a pilot channel. A transmission signal transmitted by a transmission side, or a Node B, is expressed asTx=WdAdSd+WpApSp  (1)
In equation (1), Wi is a Walsh code which is a spreading code, so Wd represents a Walsh code used for a traffic channel and Wp represents a Walsh code used for a pilot channel. Further, in Equation (1), Ai is a channel gain, so Ad represents a channel gain of a traffic channel and Ap represents a channel gain of a pilot channel. Moreover, in Equation (1), Si represents each of symbols constituting a packet as mentioned above, Sd represents a symbol on a traffic channel, and Sp represents a symbol on a pilot channel. However, the Sp uses a pattern previously agreed between a transmission side, or a Node B, and a reception side, or a UE.
A communication system employing HSDPA (hereinafter referred to as an “HSDPA communication system”) transmits a signal by the packet, and one packet is comprised of a plurality of time slots. A transmission unit by the packet is a transmission time interval (hereinafter referred to as “TTI”), and one TTI is comprised of 3 time slots. Further, the number of symbols transmitted for one time slot is variable according to a spreading factor (hereinafter referred to as “SF”) applied to the corresponding time slot. In the HSDPA communication system, SF=16 is generally used, so 480 symbols are transmitted for each packet. As a result, 160 symbols are transmitted for each time slot.
In 16QAM, one symbol is comprised of 4 bits, so 1920 bits are randomly generated for each packet, and in QPSK, one symbol is comprised of 2 bits, so 960 bits are randomly generated for each packet. In the case of 16QAM, when one packet is transmitted, 480 symbols are transmitted, and if the 480 symbols are evenly generated as 120 inner symbols, 240 middle symbols and 120 outer symbols, average power of the 480 symbols within one packet will become 1 (<Si>=1). However, generally, the 480 symbols within one packet are not uniformly generated as 120 inner symbols, 240 middle symbols and 120 outer symbols as stated above in view of a characteristic of data. For example, when 1920 bits constituting the 480 symbols are all generated with 0, the 480 symbols are all generated as inner symbols of A+jA on the constellation illustrated in FIG. 1. Thus, average power <Si> of the 480 symbols becomes 0.2 (<Si>=0.2). If average power <Si> of the 480 symbols is 0.2, a reception side cannot but estimate the average power <Si> as 0.2, even when there is no noise or distortion. In contrast, however, if 1920 bits constituting the 480 symbols are all generated with 1, the 480 symbols are all generated as outer symbols of 3A+3jA on the constellation illustrated in FIG. 1, so average power <Si> of the 480 symbols becomes 1.8 (<Si>=1.8). Likewise, if average power <Si> of the 480 symbols is 1.8, the reception side cannot but estimate the average power <Si> as 1.8, even though there is no noise or distortion. The uneven average power of a transmission signal, which is not 1, is called “unequal average power.”
A characteristic of the unequal average power will now be described with reference to FIG. 2.
FIG. 2 is a graph illustrating an example of a general characteristic of unequal average power when 16QAM is applied. Specifically, FIG. 2 illustrates a characteristic of a probability density function (hereinafter referred to as “PDF”) for average power of a transmission packet when 90% of transmission power is applied to a traffic channel on the assumption that the total transmission power is 1. If the 480 symbols are uniformly generated as 120 inner symbols, 240 middle symbols and 120 outer symbols during transmission of one packet, average power p of a traffic channel becomes 0.9 (P=A2d<Sd>=A2d=0.9). However, as mentioned above, there is a rare case where 480 symbols are ideally uniformly generated as 120 inner symbols, 240 middle symbols and 120 outer symbols during transmission of one packet. Generally, PDF shows a distribution characteristic with mean m=0.9 and standard deviation σ=0.0232.
If transmission power assigned to a traffic channel is 90% of the total transmission power (A2d=0.9), average power <Sd> of traffic channel symbols is not 1 but 0.9, and the traffic channel symbols are received at a reception side together with AWGN having power of 0.2 (<N>0.2), then a power ratio between a traffic channel and a pilot channel is detected in the following way by using an accumulation averaging technique of a traffic channel which is a general blind power ratio detection technique. Here, <N> represents average power of a noise. A description will now be made of a procedure for detecting a power ratio between a traffic channel and a pilot channel in the accumulation averaging technique.
If it is assumed that a channel is mixed with the AWGN, a reception side receives a signal defined asRx=WdAdSd+WpApSp+N  (2)
If only a traffic channel signal is separated from the received signal Rx of Equation (2), the separated traffic channel signal is expressed by Equation (3) below. In order to separate only a traffic channel signal from the received signal Rx, a transmission side simply multiplies the received signal Rx by the same Walsh code as a Walsh code applied to the traffic channel, for despreading.Rxd=AdSd+N  (3)
In Equation (3), Rxd is a received signal for which only a traffic channel signal is considered. In order to calculate a channel gain Ad applied to the traffic channel, accumulated average power is calculated byP=A2d<Sd>+<N>  (4)
In Equation (4), P represents accumulated average power, i.e., accumulated average power of a traffic channel. If it is assumed in Equation (4) that <Sd>=1 and <N>=0, i.e., if average power and noise power of symbols within a packet transmitted over a traffic channel are 1 and 0, respectively, then the accumulated average power can be detected as P=A2d=0.9. However, if <Sd>=0.9 and <N>=0.2 as assumed above, P=A2d<Sd>+<N>=1.01. In this case, P≠A2d, so it is not possible to detect correct A2d.
A general structure of a receiver in a mobile communication system will now be described with reference to FIG. 3.
FIG. 3 is a block diagram illustrating an example of a general structure of a receiver in a mobile communication system. Referring to FIG. 3, a reception signal Rx received at the receiver after passing a fading channel, i.e., after suffering a fading phenomenon, can be defined asRx=α·(WdAdSd+WpApSp)e−jθ+N)  (5)
In Equation (5), αe−jθ represents distortion of amplitude and phase due to a fading channel. Specifically, α represents amplitude distortion, and e−jθ represents phase distortion. The other components in Equation (5) are equal to those described in conjunction with Equation (1).
The reception signal Rx expressed by Equation (5) is applied to a despreader 310, and the despreader 310 despreads the reception signal Rx with a predetermined spreading code to separate the reception signal Rx into a traffic channel signal and a pilot channel signal, and provides the traffic channel signal to a channel compensator 320 and the pilot channel signal to a channel estimator 330. That is, the despreader 310 despreads the reception signal Rx using the same spreading code as a spreading code applied to a traffic channel in a transmitter to separate a traffic channel signal from the reception signal Rx, and provides the traffic channel signal to the channel compensator 320. Further, the despreader 310 despreads the reception signal Rx using the same spreading code as a spreading code applied to a pilot channel in the transmitter to separate a pilot channel signal from the reception signal Rx, and provides the pilot channel signal to the channel estimator 330. The traffic channel signal output from the despreader 310 is represented by αAdSde−jθ+N, and the pilot channel signal output from the despreader 310 is represented by αApSpe−jθ+N.
Meanwhile, the channel estimator 330, when it operates ideally, detects Apαe−jθ by multiplying the pilot channel signal by a complex conjugate Sp*=1−j of a pilot symbol Sp=1+j previously agreed upon between the transmitter and a receiver and normalizing the multiplication result, and then outputs a complex conjugate value of a fading channel, and the complex conjugate value is represented by(Apαe−jθ)*  (6)
As a result, the signal (Apαe−jθ)* output from the channel estimator 330 becomes an estimation value of a pilot channel for which a fading phenomenon was considered. The channel estimator 330 provides the (Apαe−jθ)* to the channel compensator 320 and a power ratio detector 340. The power ratio detector 340 serves as a traffic-versus-pilot channel power ratio detector for detecting a power ratio between a traffic channel and a pilot channel.
The channel compensator 320 performs channel compensation on the traffic channel by using the (Apαe−jθ)* output from the channel estimator 330, and the channel-compensated traffic channel signal is expressed asαAdSde−jθ+N×(Apαe−jθ)*=|α|2AdApSd+N′  (7)
The channel compensator 320 generates a channel compensation signal |α|2AdApSd+N′ by multiplying the despread traffic channel signal αAdSde−jθ+N by the channel estimation signal (Apαe−jθ)* output from the channel estimator 330, and provides the generated channel compensation signal |α|2AdApSd+N′ to the power ratio detector 340. That is, the channel compensation signal |α|2AdApSd+N′ output from the channel compensator 320 becomes a signal phase-compensated multiplying the traffic channel signal αAdSde−jθ+N output from the despreader 310 by the channel estimation signal (Apαe−jθ)* output from the channel estimator 330. The power ratio detector 340 detects a power ratio between a traffic channel and a pilot channel by using the channel-compensated signal |α|2AdApSd+N′ output from the channel compensator 320 and the channel estimation signal (Apαe−jθ)* output from the channel estimator 330.
An operation of the power ratio detector 340 will now be described herein below.
The power ratio detector 340 first detects accumulated average power of the channel compensation signal |α|2AdApSd+N′ output from the channel compensator 320. The accumulated average power for the channel compensation signal |α|2AdApSd+N′ is defined as|α|4(AdAp)2<Sd>+<N′>  (8)
In Equation (8), in an ideal case, <Sd>=1 and <N′>=0, so accurate |α|4(AdAp)2 can be detected. However, in an actual radio channel environment, <Sd>≠1 and <N′>≠0, so Equation (8) can rewritten as |α|4(AdAp)2  (9)
The accumulated average power is expressed as |α|4(AdAp)2 in Equation (9) since <Sd>≠1 and <N′>≠0, it becomes a value different from the accumulated average power |α|4(AdAp)2 in the ideal case. A square root of the accumulated average power represented by Equation (9) is expressed as√{square root over ( |α|4(AdAp))}= |α|2(AdAp)  (10)
The power ratio detector 340 detects a power ratio between a traffic channel and a pilot channel by calculating the square root of the accumulated average power shown in Equation (10) as a square of the channel estimation signal (Apαe−jθ)* output from the channel estimator 330, and this can be expressed as
                                                                                                                                                          α                                                              4                                    ⁢                                                            (                                                                        A                          d                                                ⁢                                                  A                          p                                                                    )                                        2                                                  <                                  S                  d                                >                                  +                                      <                                          N                      ′                                        >                                                              _                                                                                          α                                            2                        ⁢                          A              p              2                                      =                                                                                                                      α                                                        2                                ⁢                                  (                                                            A                      d                                        ⁢                                          A                      p                                                        )                                            _                                                                                          α                                                  2                            ⁢                              A                p                2                                              =                                    A              d                                      A              p                                                          (        11        )            
In Equation (11), if <Sd>≠1 and <N′>≠0, the output of the power ratio detector 340 includes not only the power ratio
      A    d        A    p  between a traffic channel and a pilot channel but also an error component. In addition, since <Sd>≠1, it will be assumed that <Sd>1+Δ<Sd>. Then, the output of the power ratio detector 340 is expressed as
                                                                                                                                                          α                                                              4                                    ⁢                                                            (                                                                        A                          d                                                ⁢                                                  A                          p                                                                    )                                        2                                                  <                                  S                  d                                >                                  +                                      <                                          N                      ′                                        >                                                              _                                                                                          α                                            2                        ⁢                          A              p              2                                      =                                                            (                                                      A                    d                                                        A                    p                                                  )                            2                        +            error                                              (        12        )            
In Equation (12), an error component is
                    Δ        <                  S          d                >                              ·                                          (                                                      A                    d                                                        A                    p                                                  )                            2                                +                                                    <                                  N                  ′                                >                                                                                                      α                                                        4                                ⁢                                  A                  p                  4                                                      .                                                          
Meanwhile, a demodulator 350 receives a channel compensation signal |α|2AdApSd+N′ output from the channel compensator 320 and rearranges the channel compensation signal |α|2AdApSd+N′ as shown in Equation (13) below.
                                                                                        α                                            2                        ⁢                          A              d                        ⁢                          A              p                        ⁢                          S              d                                +                      N            ′                          =                                                            (                                                                                                  α                                                              2                                    ⁢                                      A                    p                    2                                                  )                            ·                              (                                                      A                    d                                                        A                    p                                                  )                                      ⁢                          S              d                                +                      N            ′                                              (        13        )            
Before actually demodulating a traffic channel signal, the demodulator 350 separates the channel compensation signal output from the channel compensator 320 into a traffic channel signal and a noise component by dividing the channel compensation signal by a signal output from the power ratio detector 340, and this can be expressed as
                                                                                          (                                                                                                            α                                                                    2                                        ⁢                                          A                      p                      2                                                        )                                ·                                  (                                                            A                      d                                                              A                      p                                                        )                                            ⁢                              S                d                                      +                          N              ′                                                          (                                                                                        α                                                        2                                ⁢                                  A                  p                  2                                            )                        ·                          (                                                A                  d                                                  A                  p                                            )                                      =                                            S              d                        +                                          N                ′                                                              (                                                                                                            α                                                                    2                                        ⁢                                          A                      p                      2                                                        )                                ·                                  (                                                            A                      d                                                              A                      p                                                        )                                                              =                                    S              d                        +                          N              ″                                                          (        14        )            
In Equation (14), N″ is a noise component.
Then, the demodulator 350 demodulates the signal of Equation (14) by the bit by using the constellation described in conjunction with FIG. 1, and outputs the demodulation result to a turbo decoder 360. The turbo decoder 360 decodes an output signal of the demodulator 350 in a turbo decoding scheme corresponding to a turbo encoding scheme applied in the transmitter, and outputs its original information bits.
As described above, when power of a noise mixed in a received signal fails to be removed, the general blind power ratio detection technique, especially the blind power ratio detection technique based on the accumulation averaging technique has difficulty in performing accurate blind power ratio detection due to the noise power. That is, since a noise component is included in the signal output from the power ratio detector 340 as a power component as described in conjunction with Equation (12), it is difficult to remove the noise component. In addition, the accumulation averaging technique can be directly affected by the unequal average power problem and is sensitive to a fading phenomenon, making it difficult to perform blind power ratio detection. When a signal is transmitted using a high-order modulation scheme in an HSDPA communication system, the general blind power ratio detection technique, especially the accumulation averaging technique has difficulty in modulating the transmitted signal.